1. Field of the Invention
The invention concerns converter circuits which convert a single ended voltage to a differential output. More particularly, the invention concerns reducing circuit complexity and size of such a converter circuit while providing temperature compensation. The invention also concerns a variable gain amplifier ("VGA") with such a converter circuit.
2. Description of the Prior Art
Many communication applications require some sort of variable gain that is a exponentially proportional to an input control voltage. Since on a dB scale the gain curve becomes a straight line, this is commonly referred to as "linear-in-dB". An example of where a linear-in-dB variable gain is required is in transceivers for cellular phones. A VGA is used in the automatic gain control loop of the transmitter to regulate the power of the signal transmitted from the cellular phone. A VGA is also used in the receiver to regulate the signal power for the intermediate-frequency (IF) and signal dividing stages of the receiver despite a varying input power of the received RF signal.
Most VGA circuits accomplish this desired linear-in-dB behavior by, one way or another, exploiting the exponential characteristic of a bipolar transistor. A well-known technique relies on the fact that the ratio of the collector currents of a bipolar differential pair is exponentially dependant on the differential input voltage. FIG. 1 shows a differential pair of bipolar transistors Q1 and Q2 with common emitters biased by a tail current Itail from current source 5 and their bases controlled by a differential input voltage vin+, vin- at differential inputs 1,2. The relation between the collector currents iout+ and iout- at the outputs 3, 4 and the input voltages vin+ and vin- can be described as ##EQU1## where q is the charge of an electron, k is Boltzmann's constant and T represents the absolute temperature. The fact that it is the ratio of the collector currents that exhibits the linear-in-dB behavior enables the bipolar differential pair to be used for a wide range of variable gain circuits that rely on current ratios to set their gain. A classic and widely-used example of such a circuit is the translinear Gilbert multiplier cell known, inter alia, from, B. Gilbert, "Analog IC Design, the Current Mode Approach", Chapter 2, Peter Peregrinus Ltd (U.K. 1990).
Equation 1 clearly shows the exponential characteristic of the circuit. It also reveals another important aspect. The presence of the absolute temperature T, in the denominator of the right hand side argument, indicates that the collector current ratio is not only a function of the differential input voltage, but also of the operating temperature. This temperature effect can be quite significant, as circuits are commonly required to operate over a temperature range between about 230.degree. K. and 380.degree. K. The mathematical solution to the temperature sensitivity of Eq. 1 is relatively simple: multiplying the differential input voltage with a factor that is proportional to the absolute temperature cancels out the absolute temperature T in the denominator. In other words, if EQU f (T)=cT (2)
where c is an arbitrary constant, then by multiplying the differential input voltage with f(T), Eq. 1 becomes ##EQU2## The right hand term of Eq. 3, apart from the constants, relies on the input voltage alone and has become independent of the temperature.
A known way to realize the temperature cancellation principle of Eq. 3 in a physical circuit is shown in FIGS. 2(a), 2(b). FIG. 2(b) represents a Gilbert multiplier that multiplies an incoming control signal by a factor that equals the ratio of the two currents Iconst and Iptat. If the current "Iconst" is constant over temperature and "Iptat" is proportional to the absolute temperature T (PTAT), this ratio becomes the desired linear function of the temperature: ##EQU3##
Unfortunately, the known multiplier of FIG. 2(b) only accepts a differential current (I0+dl, I0-dl) at its input 12, 13, whereas the required control input for most variable gain amplifiers is single-ended voltage. This disparity accounts for the added circuitry shown in FIG. 2(a), which is a schematic of a traditional voltage-to-current convertor with single-ended input and differential output. The circuit of FIG. 2(a) is known from: Gurkanwal Singh Sahota, Charles James Persico, "High Dynamic Range Variable-Gain Amplifier for CDMA Wireless Applications", proceeding ISSCC (U.S.A. 1997). To understand the operation of the circuit of FIG. 2(a), assume that the amplifier A1 has sufficient gain to keep the voltage at its positive input equal to the reference voltage V.sub.ref at its negative input. In that case, the voltage at the right hand terminal of the input resistor R1 becomes V.sub.ref. Since the other terminal of the resistor R1 is connected to the input terminal 9 which receives the gain control voltage V.sub.gain, there will be a voltage drop of V.sub.ref -V.sub.gain across the resistor R1, which causes a current dI to be taken away from the circuit. This current dI is supplied by the transistor Q3, together with the constant bias current I0. The total current at the collector of the transistor Q3 is therefore I0+dI.
In case the voltage at the positive input of the amplifier A1 inadvertently deviates from the assumed voltage V.sub.ref, the feedback loop formed by the transistors Q1, Q2 and Q3 will adjust the collector current of the transistor Q3 until the voltage at the positive input of the amplifier A1 is corrected to V.sub.ref. Due to the parallel connections of the base and emitter terminals of the transistors Q3 and Q4, the collector current of the transistor Q4 will track that of the transistor Q3. Thus, a current of I0+dI will flow out of the first output terminal 10 of the voltage-to-current converter.
The transistor Q5 also copies the collector current of the transistor Q3. In this case, however, the current I0+dI is directed through the current mirror formed by the transistors Q6/Q7, and then subtracted from a constant bias current 2I0. The result at the second output terminal 11 is a current that equals I0-dI. The total differential output current of the circuit in FIG. 2(a) is ((I0+dI-(I0-dI)), or 2dI, which can be used to directly drive the Gilbert multiplier of FIG. 2(b).
Looking more closely at the voltage-to-current converter of FIG. 2(a), several drawbacks become apparent, most of them related to the accuracy of the circuit. First, the voltages at the collectors of the transistors Q2, Q3, Q4 and Q5 are all different, leading to small but significant differences in the collector currents of the respective transistors. This causes an error in the overall gain setting of the variable gain amplifier. In the same way, integrated circuit process related random mismatches between any of the devices Q2-Q5 will adversely affect the accuracy. A second drawback stems from the fact that the current coming out of the transistor Q5 is first mirrored by the transistors Q6 and Q7, while the current from the transistor Q4 is directly flowing to the output terminal 10 without first being mirrored by a current mirror. Not only will any random mismatch between the transistors Q6 and Q7 deteriorate the overall performance, but also here, the collector voltages of the two current mirror transistors Q6, Q7 are not identical, adding to the total error. A final, more general, drawback involves the complexity of the circuit. Combining the two schematics of FIG. 2 (a) and FIG. 2 (b) typically yields a block that consumes a considerable part of the total die area of a VGA.